System and method of downconversion where the received signal is downconverted

ABSTRACT

A system and method for simultaneous transmission and receival where the received signal is downconverted in a downconverting unit by using a portion of the transmitted modulated signal as LO for the downconversion. The downconversion is performed to an intermediate frequency signal having a frequency equal to the difference between the frequency of the transmitted signal and the frequency of the received signal. There is provided a common antenna for transmitting and receiving, and a duplex filter in front of the antenna to separate the transmitted signal and the received signal with the transmitter data on the transmitter side having been applied to a modulator. In front of the duplex filter on the transmitter side is provided a power amplifier. On the receiver side there is, in addition, provided a demodulator to permit withdrawal of user data for the user on the receiver side. Compensation for the modulation introduced by the signal in the downconversion unit is provided in the receiver by a device where the transmitted phase is used as a reference, delayed in a time delay unit by an appropriate delay, and where this delay unit may be estimated and updated in order to compensate for variable delay in the outer signal path.

BACKGROUND OF THE INVENTION

The present invention relates to a system for wireless data transmittersand receivers of the type disclosed below, and to a method fordownconversion using the modulated signal on the transmitter side.

More specifically, the present invention relates to wireless datatransmitters and receivers, where transmission and reception occur atthe same time, but at different frequencies, i.e., FDM (frequencydivision multiplex) to achieve full duplex data communication.

DESCRIPTION OF THE RELATED ART

In systems for wireless (radio) data communication there are mechanismsfor allocation of frequency slots for the transmission and receiverchannels. This access mechanism is denoted FDMA (frequency divisionmultiple access). In some systems there is a constant difference in thecarrier frequency between a transmitted and received signal. One exampleof such a system is Inmarsat Standard M. In this case, it is sufficientto have one frequency synthesizer which is capable of serving both thetransmitter and the receiver. By using this system cost savings may beachieved.

SUMMARY OF THE INVENTION

According to the present invention, further savings are possible if thetransmitted signal itself is used for downconversion of the receivedsignal.

Furthermore, it is an objective of the invention to connect it tomodulation methods of the constant envelope type, such as CPM(continuous phase modulation), DPM (digital phase modulation) or nearconstant envelope modulation such as, e.g., the OQPSK type (offsetquadrature phase shift keying).

In U.S. Pat. No. 5,444,737 only FM modulation or FSK modulation isdescribed. The present invention is more general and is capable ofmodelling all types of constant envelope modulation methods which may bedescribed as either digital phase modulation (DPM) or digital frequencymodulation (CPM). The present invention may be used for the OQPSKmodulation method (offset quadrature amplitude modulation): filteredOQPSK is, at the outset, not constant envelope modulation, but byelimination of the amplitude portion (hardlimiting) in the transmitter,the transmitted signal will become a constant envelope signal. Thismeans that the amplitude variation is eliminated. OQPSK has thecharacteristic that the losses associated with such hardlimiting arenegligible. With the present invention there are described allmodulation methods which either per se possess constant envelope orwhere constant envelope is provided by hardlimiting in the transmittereither by digital signal processing where the amplitude portion issuppressed, or by a hardlimiting output amplifier (HPA).

In the aforementioned US patent an analog data signal is used which istaken from the transmitter section and which is subtracted from thereceived signal after a frequency discriminator and an appropriateanalog delay.

With the present invention it is not necessary to use a frequencydiscriminator, but the phase portion (the amplitude is neglected) of thetransmitted signal is derived directly from the transmitter signal, thensign inverting is carried out, and from this is formed by exponentiationa complex equivalent baseband signal. By multiplying the receivedcomplex equivalent baseband signal in the receiver by the complexequivalent baseband signal for the aforementioned derivative of thetransmitted signal, there will be compensation for the unwanted phaseperturbation which occurs as a result of downconversion with themodulated transmitted signal used as a local oscillator derived from thetransmission signal.

In the aforementioned US patent there is used an analog embodiment whichcan prove to be rather impractical. The present invention prescribes afully digital embodiment by use of, e.g., a DSP (digital signalprocessor).

In the US patent a fixed delay is utilized in an analog execution withRC elements to compensate for the delay in the TX/LO loop. This is notvery practical, and with the present invention this is carried outdigitally with the aid of a digital Lagrange interpolating filter. Usingan LMS algorithm (LMS=least mean square) the delay is either calculatedduring the production phase, thereafter being set as a fixed value inthe radio transceiver, or the delay may be calculated continuouslyduring operation of the radio transceiver with the same as algorithm(adaptive algorithm). In the first case, the LMS algorithm is a part ofthe production set-up, and in the latter case this algorithm is a partof the radio equipment. This method provides for increased flexibilityand a considerably more accurate compensation of the modulation of thesignal derived from the transmitter in the event of componentvariations, temperature drift and aging which may occur in the outersignal path, in contrast to what is previously known.

The aforementioned effects are brought about with the aid of a system ofthe type introduced above, the characteristic features of which are setforth below, and a method of the type introduced above, thecharacteristic features of which are set forth below.

BRIEF DESCRIPTION OF THE DRAWINGS

The invention will be described in more detail in the following, withreference to the drawings wherein:

FIG. 1 is a block diagram of a point to multipoint full duplex radiocommunication system.

FIG. 2 is a detailed block diagram of the system in accordance with thepresent invention.

FIG. 3 shows the LMS update of the delay.

FIG. 4 shows the Lagrange interpolator.

FIG. 5 shows an alternative downconversion structure.

DESCRIPTION OF THE PREFERRED EMBODIMENTS

The radio transceiver, shown to the right in the block diagram in FIG.1, uses a common antenna 20 to transmit (TX) and to receive (RX). Thecarrier frequency of the transmitted signal is controlled digitally by areference signal 37 fed into a phase locked loop (PLL) 36, via adigital-to-analog converter (DAC) and a device for direct digitalfrequency synthesis (DDS) 40, cf. FIG. 2. By adding a data signal 46 b,the latter derived in the preceding differentiator 44, to a referencefrequency 42, the data sequence will yield a modulated signal at theoutput of the modulator 33, cf. FIG. 1.

With the aid of a coupler 32, a suitable portion 35 of the transmittedsignal can be used as a local oscillator (LO) for the receiver signal22. The receiver signal is downconverted to an intermediate frequency(IF), which is the difference between the transmitter's carrier wavefrequency f₂ and the incoming signal's carrier wave frequency f₁. Thephase of the received IF signal 26 is perturbed by the modulation in thetransmitted signal. Compensation for this perturbation is achieved byusing the transmitter's baseband signal 46 a. The latter signal isconverted to a complex vector representation 81 by exponentiation 82,followed by multiplication 80 of this complex vector by the incomingsignal's complex equivalent baseband representation 76. Both the phase46 a of the transmitter signal and the receiver signal 76 arerepresented digitally, and the compensation is performed fullydigitally.

By the term “the outer signal path” is meant the signal path from thetransmitter through the transmitter's PLL 36, the coupler 32, thedownconverter 25 and the demodulator 27, cf. FIG. 1 and FIG. 2. By theterm “the inner signal path” is meant the signal path through thevariable delay 53, the sign inverter 86 and the unit which implementsthe exponentiation 82. Owing to delay and filtering in the outer signalpath, appropriate delay 53 must be introduced in the compensation branchin order to eliminate completely the aforementioned phase errorintroduced by the transmitted signal employed as LO. The delay in theouter signal path is variable as a result of spread in the componentdata, aging and temperature drift. A continuous estimation of this delayand correspondingly an adjustment of the delay in the delay unit 53 ofthe inner signal path will therefore improve the performance of thereceiver considerably. The present invention describes two alternativemethods of implementing this delay. In the first alternative, the delayis estimated and adjusted during the production phase in order to hold afixed value thereafter, and in the second alternative the delay isestimated and adjusted continuously during use. There are alsoprescribed two alternatives for downconversion. In the first alternativethe difference between the transmitter and receiver frequency comprisesthe intermediate frequency for the receiver. In the second alternativethe portion of the transmitter signal 35 which is used as the receiver'sLO will first be downconverted at a frequency corresponding to thedifference between the frequencies f₂ and f₁. After filtering out thelower sideband of the mixer product which emerges after the firstdownconversion, the latter mixer product is used as LO for a quadraturedemodulator 60 for direct downconversion of the received signal 24 to abaseband signal given by the signals 63 and 64.

The present invention is applicable both for point to pointcommunication and for point to multipoint communication, such as datacommunication between a user terminal and a base station in a mobilecellular system or a land earth station (LES) in a satellitecommunication system.

The invention will be described in more detail below with reference tothe figures, wherein FIG. 1, as mentioned above, shows a radiocommunication system for full duplex radio communication. Here there isshown a point to multipoint system common in most mobile communicationsystems. In an earthbound, cellular system, this means communicationbetween the base stations and several terminals. In FIG. 1 only oneterminal is shown. In satellite communication point to multipointcommunication means communication between a satellite earth station(LES=land earth station) via one or more satellites to a plurality ofsatellite terminals.

A common antenna 20 is used for both transmission and receival. A duplexfilter 21 in front of the antenna separates the transmitted signal 29from the received signal 22, cf. FIG. 1. On the transmitter side thetransmitted data 34 are applied to a modulator 33. The coupler 32couples a portion 35 of the modulated transmitter signal 31 to thereceiver's downconversion unit, mixer 25. Transmitter signal 31 is alsosent further to the output amplifier (HPA) 30 in front of duplex filter21. On the receiver side a low noise amplifier (LNA) 23 amplifies theincoming signal, whereafter this signal is downconverted to anintermediate frequency as explained above. The intermediate frequency,as mentioned previously, is equal to the difference between thetransmitted carrier wave frequency f₂ and the received carrier wavefrequency f₁. After the demodulator 27, the received data 28 appear.

FIG. 2 shows a detailed block diagram of the invention. The descriptionof this block diagram follows the signal path of the transmitted datafrom the input 34 to the antenna 20, including the signal processing onthe transmitter side, and the received signal from the antenna 20 to thereceived data 28, including the signal processing in the receiversection.

The transmitted data 34 are converted to a complex signal 51 in a realto complex converter 52. In the case of 4 PSK modulation, this meansthat the incoming bit is alternately applied to the real part and theimaginary part. To obtain an oversampling factor N of the transmitteddata signal, an interpolation filter 50 is provided. The transmittedsignals are then given an appropriate pulse shape in a pulse shapingfilter 48. In order to form a constant envelope signal from the filteredtransmitter signal 47, the argument operator 45 takes the argument (thephase) φ(iT/N) 46 a (i is the count variable for time, T/N is thedifference in time between consequtive samples) for the complex signal47. The phase signal φ(iT/N) 46 a is essential for the presentinvention. As is apparent from the block diagrams in FIG. 1 and FIG. 2,this signal is used to compensate for the previously mentioned phaseerror introduced by signal 35 during the downconversion in mixer 25.After the argument operator 45 (block diagram in FIG. 2), the signal isdifferentiated in 44 so that the instantaneous frequency deviation froma carrier wave 46 b emerges. To this frequency deviation is added thefrequency f_(c)/M of the carrier wave. M emerges as a result of the factthat an upmultiplication of the frequency corresponding to M is carriedout in the following phase locked loop (PLL) 36. The resultant frequencysignal 41 is then applied to a DDS (direct digital synthesizer) 40 and aDAC (digital to analog converter) 40 and to a “smoothing” oranti-aliasing filter 38, which generates an analog phase modulatedsignal 37 as a reference signal for the upmultiplying VCO-controlledphase locked loop 36 (VCO=voltage controlled oscillator). The endprocessing of the signal in the radio section and the receival furtheron to the IF filter 58 is described above and will not be repeated here.

The further signal processing in the demodulator is described in thefollowing. The output from the IF filter 58 is downconverted in aquadrature mixer which separates the cosine and sine part 63 and 64 ofthe received signal. After necessary filtering and amplification 65 and66, the two components are converted 70 and 71 to digitally representedsignals 73 and 74. The conversion rate N/T is the same as in thetransmitter unit. The cosine part is converted to the real part and thesine part is made the imaginary part in the following unit 75, yieldinga complex equivalent baseband signal 76. The latter signal is multipliedin the multiplier 80 by the previously mentioned complex compensationsignal 81, and this constitutes the core of the invention. Thismultiplication implements a phase correction consisting of threecomponents:

a) phase error for the received signal 79

b) frequency error for the received signal

c) compensation for phase error introduced when portion 35 of thetransmitted signal is used as LO for downconversion.

As indicated in FIG. 2 estimation of phase and frequency error isimplemented with the aid of decision directed techniques, which areknown per se techniques, but the present invention combines theaforementioned compensation with correction of phase-frequency error ina simple and effective manner.

In the following is a description of the compensation mentioned in c)above. The reference for the compensation is taken from the transmittersphase signal 46 a. This signal φ(iT/N) is delayed in a variable delayunit 53. The delay is estimated by using a derivation of the well-knownLMS algorithm, a derivation developed particularly for this purpose. Theestimated delay is denoted by {circumflex over (τ)} and is an estimateof the delay in the outer signal path. The delay unit is implemented asLagrange interpolator, cf. FIG. 4. The Lagrange in terpolator enablesrealization of an arbitrary delay for a time discrete (sampled) signal.The delayed signal may be expressed by the equation:${\hat{\phi}\left( {i\frac{T}{N}} \right)} = {\phi \left( {{i\frac{T}{N}} - \hat{\tau}} \right)}$

This signal is sign inverted 86 and added to the phase and frequencyerror signals 79 to form the composite phase correction signal$\left( {i\frac{T}{N}} \right)$

83 before the latter is calculated by exponentiation 82 and multipliedby the incoming baseband signal 76. In order to minimize white noise andintersymbol interference, an appropriate signal matched filter 88 isinserted in front of the detection unit 90.

The LSM algorithm uses:

a) an error signal 57 from the detection unit 90,

b) the input signal 87 to the matched filter 88,

c) the output signal 54 from the variable delay unit 53

in order to make an update 53 a of estimated delay {circumflex over(τ)}.

In the following the LMS algorithm is described with reference to theblock diagram in FIG. 3, and the description assumes a fully digitalimplementation of the algorithm using, e.g., a DSP (digital signalprocessor).

The estimate of the delay {circumflex over (τ)} at time (i+1)T/N isbased on an earlier estimate of {circumflex over (τ)} at time iT/Naccording to the algorithm $\begin{matrix}{{\hat{\tau}\left\lbrack {\left( {i + 1} \right)\frac{T}{N}} \right\rbrack} = {{\hat{\tau}\left( {i\frac{T}{N}} \right)} - {\frac{1}{2}\beta {\nabla\left( {i\frac{T}{N}} \right)}}}} & (1)\end{matrix}$

where β is the update constant and {overscore (V)}(iT/N) is the gradientwith respect to {circumflex over (τ)} defined by $\begin{matrix}{{\nabla\left( {i\frac{T}{N}} \right)} = \frac{\partial{J\left( {i\frac{T}{N}} \right)}}{\partial{\hat{\tau}\left( {i\frac{T}{N}} \right)}}} & (2)\end{matrix}$

where J is the mean square error at time iT/N. In the following iT/N isreplaced by i in order to simplify the terminology in the furtherdescription. By using the equations (1) and (2), it can be shown thatthe update algorithm will be

{circumflex over (τ)}(i+1)={circumflex over(τ)}(i)+βsign{image[e(i)·d(i)]  (3)

where $\begin{matrix}{{d(i)} = {\left\lbrack {\frac{\partial{\hat{\phi}(i)}}{\partial t} \cdot {r(i)}} \right\rbrack*{m(i)}}} & (4)\end{matrix}$

In equation (4) the operator * denotes convolution. For the furtherdescription, the following signals are defined as indicated below:

{circumflex over (τ)}(i) the estimate of the delay 111 at time i (cf.FIG. 4)

{circumflex over (τ)}(i+1) the updated estimate 53 a of the delay attime i+1

e(i) the error signal 57 at the output of the detection unit

d(i) the output signal 101 from the signal matched filter 100

r(i) the received signal 56 at the input to the multiplier 99

{circumflex over (φ)}(i) the estimated phase 54 at the output of thevariable delay unit 53

m(i) the impulse response of the matched filters 88 and 100

The output signal {circumflex over (φ)}(i) 54 from the variable delayunit 53 is differentiated in the differentiator 97, the output signal 98is then multiplied by the received signal r(i) 56, and the result isconvolved with the matched filter 100 with impulse response m(i). Theoutput d(i) 101 from the matched filter 100 is then multiplied 102 bythe error signal e(i) 57, and the imaginary part 104 is taken therefrom.The result 103 is an estimate of the gradient. A simplification of theLMS algorithm is the use of the sign for the gradient 106 which formsstill another estimate of the gradient 107 when this is multiplied by anappropriate update constant β. The integrator 109 thus yields an updatedvalue of the estimated delay in the delay unit 53.

As described above, the present invention utilizes a derivation of theLMS algorithm developed particularly for this purpose. With the presentinvention it is thus possible to adjust for delay variations in theouter signal path.

The variable delay is obtained using a fourth order Lagrangeinterpolator realized as an FIR (finite impulse response) filter wherethe taps h₀, h₁, h₂, h₃ in the filter are calculated for each updateaccording to the algorithm displayed in FIG. 4.

The present invention may be utilized in an alternative downconversionstructure described in the introduction, where the intermediatefrequency stage is omitted and direct conversion to baseband takesplace, as shown in FIG. 5.

What is claimed is:
 1. A system for simultaneous transmission andreception where a received signal is downconverted, by using atransmitted signal (35), employed as a local oscillator (LO),characterized in that a digital signal processor (DSP) is provided whichcompensates for unwanted modulation in the transmitted signal (35) usedfor downconversion, that a digital complex baseband reference signal (46a) is used for elimination of the unwanted modulation in the transmittedsignal where a part (35) of the transmitted signal is employed as thelocal oscillator (LO), said elimination being performed digitally withthe aid of a digital signal processing algorithm in complex baseband,and that there is provided a multiplier (80) for said elimination, therebeing carried out an inverse operation of the multiplication in themultiplier (25) which relates to every kind of phase and/or frequencymodulation in the local oscillator (LO) to which the received signal(24) is applied.
 2. The system according to claim 1, characterized inthat for the downconversion there may be provided devices fordownconversion in two stages, first at a frequency equal to thedifference (f₂−f₁) between transmitted and received carrier waves, andthen a second stage where the received signal is downconverted directlyto baseband with the aid of a quadrature downconverter.
 3. The systemaccording to claims 1, characterized in that a digital signal processor(DSP) is provided for compensation of the unwanted modulation with theaid of digital signal processing algorithms.
 4. The system according toclaims 1, characterized in that for elimination of said unwantedmodulation in the LO signal (35) there is provided a delay unit (53),which delays the reference phase signal (46 a), which is representeddigitally and time discretely, a sign inverter (86) where the signal issign inverted and added to a correction signal for phase and frequencyerror in the received signal, an exponentiation unit (82) where theresultant signal is exponentiated and a multiplier wherein amultiplication by an incoming received baseband signal (76) is carriedout.
 5. The system according to claims 1, characterized in that thedelay is given an arbitrary resolution by digital calculation with theaid of a known per se Lagrange interpolation algorithm.
 6. The systemaccording to claims 1, characterized in that for estimation of optimaltime delay (53) there is utilized a special mathematical derivation ofthe known per se LMS algorithm developed particularly for this purpose.7. The system according to claims 1, characterized in that for derivingthe LMS algorithm for an updated delay (53 a) the time derivative forthe delayed phase (54 and 55) is obtained in a digital differentiator(97) and the output signal (98) is multiplied in a multiplier (99) bythe received baseband signal (87 and 56), filtered in a matched filter(100), multiplied by an error signal (57) derived from a detection unit(94), followed by taking the imaginary part (104) and the sign (106)thereof, multiplied by an appropriate constant in a multiplier (108), inorder to find an update of the delay, which integration (109) yields anupdated delay (53 a).
 8. The system according to claims 1, characterizedin that the LMS algorithm for estimation of optimal delay is appliedeither by training of the algorithm in the course of the productiontest, with the delay being fixed thereafter, whereby the LMS algorithmfor estimation of optimal delay is formed as a part of the testequipment, or the algorithm is trained by using real data in operationof the radio transceiver, in which case the LMS algorithm is a part ofthe radio transceiver.
 9. A method for simultaneous transmission andreception where a received signal is downconverted, using a transmittedsignal (35), employed as a local oscillator (LO), characterized in thata digital signal processor (DSP) compensates for unwanted modulation inthe transmitted signal (35) used for downconversion, that a digitalcomplex baseband reference signal (46 a) is used for elimination of theunwanted modulation in the transmitted signal where a part of thetransmitted signal is employed as the local oscillator (LO), saidelimination being performed digitally with the aid of a digital signalprocessing algorithm in complex baseband, and that a multiplier (80)provides for said elimination, there being carried out an inverseoperation of the multiplication in the multiplier (25) which relates toevery kind of phase and/or frequency modulation in the local oscillator(LO) to which the received signal (24) is applied.
 10. The methodaccording to claim 9, characterized in that a downconversion is carriedout in two stages, first at a frequency equal to the difference (f₂−f₁)between transmitted and received carrier wave, and then a second stagewhere the received signal is downconverted directly to baseband with theaid of a quadrature downconverter.
 11. The method according to claims 9,characterized in that a digital signal processor (DSP) compensates forthe unwanted modulation with the aid of digital signal processingalgorithms.
 12. The method according to claims 9, characterized in thatsaid unwanted modulation in the LO signal (35) is eliminated in that thereference phase signal (46 a), which is represented digitally and timediscretely, is delayed in a delay unit (53), sign inverted in a signinverter (86) where the sign inverted signal (85) is added to acorrection signal for phase and frequency error in the received basebandsignal, the resultant signal (35) is exponentiated in an exponentiationunit (82) and multiplied in a multiplier by the incoming receivedbaseband signal (76).
 13. The method according to claims 9,characterized in that the delay is given an arbitrary resolution bydigital calculation with the aid of a known per se Lagrangeinterpolation algorithm.
 14. The method according to claims 9,characterized in that to estimate optimal time delay (53), a specialderivation of the known per se LMS algorithm developed for this purposeis used.
 15. The method according to claims 9, characterized in that fordeveloping the LMS algorithm for updated delay (53 a), the timederivative for the delayed phase (54 and 55) is obtained in a digitaldifferentiator (97) and the output signal (98) is multiplied in amultiplier (99) by the received baseband signal (87 and 56), filtered ina matched filter (100), multiplied by an error signal (57) derived froma detection unit (94), followed by taking the imaginary part (104) andthe sign (106) thereof, multipled by an appropriate constant in amultiplier (108), in order to find an update of the delay, whichintegration (109) yields an updated delay (53 a).
 16. The methodaccording to claims 9, characterized in that for the modulation, thereis employed an angle modulation, either frequency modulation, or phasemodulation, or any other modulation where the amplitude variations havebeen suppressed by applying the argument operation (45).